Near range obstacle detection and ranging aid

ABSTRACT

A monostatic frequency modulator continuous wave radar for detecting and determining the distance to near range objects. A temperature compensated one port Z-network is provided to generate a voltage of equal magnitude and opposite sign to that of the voltage reflected from the antenna.

This application is a continuation of application Ser. No. 07/919,833,filed on Jul. 27, 1992, now U.S. Pat. No. 5,359,331, which is acontinuation-in-part of application Ser. No. 07/553,890, filed on Jul.13, 1990, now U.S. Pat. No. 5,134,411.

BACKGROUND OF THE INVENTION

This invention relates generally to obstacle detection and rangingsystems and more specifically to near range obstacle detection systems.The following are a few of the applications in which such a near rangeobstacle detection system may be used:

Vehicular obstacle detection and headway control

Autonomous tele-operated vehicle obstacle detection

Space robotics

Control of work platforms and forklifts

Terrain mapping through vegetation

Weapons fusing

Battlefield surveillance

Tank gauging (determining the amount of a substance stored in acontainer)

Marine vessel docking and guidance

Airplane auto-docking

Personnel bridge docking

Airport runway incursion

Altimeter

Presence sensor for traffic light control

Ice thickness measurement

Pavement thickness measurement

Buried object detection

Underground tunnel or void detection

Perimeter security surveillance

Aid to the handicapped

Some prior art near obstacle detection systems utilize infrared andultrasonic radiation. These systems generally have disadvantages thatdiscourage their use.

Microwave radiation on the other hand is commonly used in a variety offorms of radar systems, and the advantages of microwave radar technologymake it attractive for near obstacle detection systems as well. See thearticle "Automotive Radar: A Brief Review" by D. M. Grimes and T. O.Jones in Proceedings of the IEEE, June, 1974, pp. 804-822 and therelevant prior art literature cited therein.

An important microwave operating band assigned for radar use, generallydesignated as X-band, covers the frequency range from 8.2 to 12.4 GHz.In this frequency range, microwave components are reasonable in bothsize and cost. For example, the dimensions of an X-band planar or patchantenna, suitable for near obstacle detection, are approximately 1inch×2 inches. A portion of X-band set aside by the FederalCommunications Commission for unlicensed use covers the frequency rangefrom 10.50 to 10.55 GHz. However, this limited bandwidth makes itdifficult to achieve adequate resolution for nearby targets. Forexample, with a conventional frequency modulated-continuous wave (FM-CW)radar system operating over a 50 MHz bandwidth, the minimum resolutionis approximately 10 feet, whereas a resolution of the order of inches isconsidered necessary for near obstacle detection such as vehicularwarning systems.

Where antenna mounting space is at a premium, it is possible to utilizeone antenna for both transmitting and receiving. This one antenna systemis called a monostatic radar system. A major drawback of monostaticsystems is the unwanted presence of an internally reflected signal fromthe antenna. Since the internally reflected signal may be an order ofmagnitude larger than the reflected signal from the target, the accuratedetection of a target may not be possible in a narrow band system sincethe receiver detects the composite signal consisting of the internallyreflected signal from the antenna as well as the reflected signal fromthe target.

An analogous situation occurs in bi-static or two antenna systems. Theunwanted signal is due to the leakage between the two antennas. However,this leakage signal is usually much smaller than the reflected antennasignal in the monostatic system. In most situations, the leakage signalcan be ignored. However, in some cases, an active two-port phaseshifter/attenuator or I-Q modulator is employed. This two-port devicegenerates a signal of equal magnitude and opposite sign to that of theleakage signal, cancelling the leakage signal.

OBJECT OF THE INVENTION

It is among the objects of this invention to provide a new and improvedsystem that can be utilized for near obstacle detection, tank/containergauging and ice thickness measurement.

Another object is to provide a new and improved microwave radar systemuseful as an obstacle detection system.

Another object is to provide a new and improved radar system for usewith limited frequency bandwidth and having a sufficiently highresolution.

Another object is to provide a new and improved radar system fordetecting the closest obstacle among multiple obstacles.

It is a further object to provide a monostatic radar system that can beutilized for near obstacle detection, tank/container gauging and icethickness measurement.

SUMMARY OF THE INVENTION

The present invention provides an improved monostatic radar system fordetecting near range obstacles and determining their distance. An activeone-port impedance-matching device (Z-network) or I-Q modulator isemployed. By means of a directional coupler, the Z-network "receives" aportion of the signal internally reflected from the antenna and, inturn, reflects that signal with a controlled amplitude and phase. Bymeans of the same directional coupler, this signal is then combined withthe internally reflected signal from the antenna as well as thereflected signal from the target. This total composite signal is fed tothe RF port of the mixer. If the reflected signal from the Z-network hasthe same amplitude and a phase difference of 180° from the internallyreflected signal from the antenna, then these signals will cancel eachother leaving only the target signal at the mixer RF port. The device"receives" the signal reflected from the antenna and reflects apre-determined portion of that signal with a predetermined phase change.The Z-network cancels the reflected signal from the antenna.

In accordance with this invention an obstacle detection apparatuscomprises: means for generating high frequency energy over a finitefrequency range and for frequency modulating over a limited range ofbandwidth such high frequency energy and for supplying saidfrequency-modulated (FM) high frequency energy to a first path providinga phase reference. A second path includes transmission and receivingsections, and the transmission section includes means for radiating thefrequency modulated high frequency energy into space in the form ofpropagating waves, while the receiving section includes means forreceiving a portion of said radiated energy after reflection from aremote object. The transmitting and receiving sections may be physicallyseparated, as in bistatic radar systems, or may have a common antennaand signal path section as in monostatic systems. The radiated energyportion acquires a phase shift related to the distance traveled by saidradiated energy and to the frequency of the radiated energy. In one ofsaid paths a means serves to phase shift frequency-modulated highfrequency energy in that one path in a certain configuration of repeatedcycles of frequency modulation to improve resolution. attainable withsaid limited range of bandwidth. In the receiving section, means forcomparing the phase of FM energy with the phase of the reflectedradiated energy serves to produce signals related to the phase-shifts ofsaid first and second paths and corresponding to the phase-shiftproduced over the distance travelled by the radiated energy portion andthe reflection thereof from the remote object. A distance signal isderived in accordance with the phase states of said phase shifter andrelated to the frequencies of said energy generating and modulatingmeans.

In accordance with an embodiment of this invention, the means forderiving a distance signal includes, means for transforming sinusoidalvoltage wave from the time domain to the frequency domain includingmeans for performing a Fourier analysis thereon. Also in accordance withan embodiment of this invention, the phase-shifting means is in thereference path.

BRIEF DESCRIPTION OF THE DRAWINGS

The foregoing and other objects of this invention as well as theinvention itself may be more fully understood from the followingdescription when read together with the accompanying drawings, in whichcorresponding parts are referenced by similar numerals throughout, andin which:

FIG. 1 is a schematic block diagram of a bi-static obstacle detectionand warning system embodying this invention.

FIG. 2 is a schematic graphical diagram of rotating vectors illustrativeof the operation of the system of FIG. 1.

FIGS. 3-A through 3-P are 16 graphs, schematic diagrams serving asexamples of signal data from the system of FIG. 1 and reconstructed inthe process of FIG. 4A.

FIGS. 4A and 4B are computer flow charts of the data processingperformed including the frequency transformation and distancecalculator.

FIG. 5 is a graphical diagram of a table of data illustrating theoperation of the distance calculator of FIG. 1.

FIGS. 6A, 6B, and 6C are idealized timing and waveform diagramsillustrating the operation of parts of the system of FIG. 1.

FIGS. 7, 8, and 9 are schematic block diagrams of modified forms of theinvention.

FIG. 10 is a schematic diagram showing zones in which targets subject tomonitoring are located in relation to an over-the-road vehicle.

FIG. 11 is a schematic diagram of a typical monostatic radar system.

FIG. 12 is a schematic diagram of a monostatic radar system embodyingthe one-port Z-matching device of this invention.

FIG. 13 is a schematic diagram of the temperature uncompensated designof the one-port Z-matching device.

FIG. 14a is the parasitic and temperature compensated design of theone-port Z-matching device.

FIG. 14b is the temperature compensated voltage generator.

FIG. 15 is the schematic diagram of a balanced monostatic radar systemembodying this invention.

FIG. 16 is the schematic diagram of a balanced monostatic radar system.

FIG. 17 is a schematic diagram of the monostatic system using anunbalanced Z-matching device.

FIG. 18 is a schematic diagram using a temperature compensated balancedZ simulator.

FIG. 19 is a schematic block diagram of a monostatic radar systemembodying this invention.

FIG. 20 is a schematic block diagram showing a typical frequencymodulated continuous wave (FM-CW) radar system.

FIG. 21 is a schematic block diagram showing a modified frequencymodulated-continuous wave (FM-CW) radar system.

DETAILED DESCRIPTION OF THE INVENTION

In the system 10 of FIG. 1 for radar ranging, a voltage-controlledoscillator 12 (VCO) generates the basic RF signal as a continuous wave(CW) with a periodic frequency modulation (FM) superimposed inaccordance with a periodic modulating voltage from a selectable source14.

The RF voltage from the VCO 12 is applied via a power splitter 16 andtransmitting RF patch 18 to a transmitting antenna 20. The latterradiates a corresponding electromagnetic wave to a remote target 22, andthe radiation reflected from the target returns to a receiving antenna24. The reflected RF signal is supplied to the RF input of a mixer 26.Another RF path 28 from the power splitter 16 supplies the FM-CW signalto a phase shifter 30, which shifts the phase of that signal in acertain periodic configuration and applies the phase-shifted signal tothe LO (local oscillator) input 32 of the mixer 26.

Voltage Controlled Oscillator--The frequency of the VCO 12 is determinedby the applied control voltages V₀, V₁, V₂, V₃, from the source 14 andcan be varied over the operating band in several ways. Among them are:

a. Continuously increasing or decreasing frequency over a fixed periodT. This is usually referred to as a linear frequency-modulatedcontinuous wave (linear FM-CW) signal. This type of frequency modulationcan be used with either an analog or digital signal processing system.

b. Step-wise increasing frequency is preferable for the digital signalprocessing approach. For optimum signal processing efficiency, thegenerated frequencies should be equally spaced.

The present best mode of the invention uses the step-wise frequencymodulation scheme. Four frequencies are generated during each period.They are spaced 12.5 MHz apart, beginning at 10.5 GHz and ending at10.5375 GHz. Each frequency state is energized for a period of 0.02msec. Therefore, each frequency sweep period has a 0.08 msec duration asshown in FIGS. 6A through 6C.

Power Splitter--The power splitter 16 is a 3-port device (e.g. aWilkinson power divider or a quadrature coupler with the isolated portterminated). The energy of the signal at the input port 17 is splitequally between output ports 27 and 34. The phase difference between thesignals at these ports must not change significantly for any of thefrequencies at which the system operates. The actual value of the phasedifference is not important to the function of the system.

Transmitting and Receiving Antennas--The signal energizing thetransmitting antenna 20 at its input port 33 is transmitted with adirectional pattern determined by the geometry of the antenna. Part ofthe signal is scattered with an undetermined pattern and amplitude froman object 22. Some of that scattered energy will appear at the receivingantenna 24 and is available at port 35. The amplitude of the receivedsignal depends on (i) the directional (gain) patterns of both antennas,(ii) the transmitting range from antenna 20 to target 22, (iii) thescattering characteristics of the target 22 and (iv) their range fromtarget 22 to receiving antenna 24.

The directional patterns of both antennas 20 and 24 and their relativelocation and orientation should be such that the magnitude of theleakage signal propagating directly between the antennas, that isbetween the points 33 and 35, should be minimized.

Mixer--The mixer 26 (or phase comparator) can be either a 2-or 3-portdevice. The latter version has been employed in the present embodiment.The 3-port device (e.g. a double balanced mixer) has reference, inputand output ports 32, 37 and 39, respectively. They are also commonlyreferred to as the LO, RF and IF ports respectively. (In the 2-portdevice, e.g., a single-balanced mixer, the LO, and RF ports use a singleport).

In this embodiment, the function of the mixer 26 is essentially that ofa "two-quadrant" phase comparator. As such, the voltage at output port39 IF is a sinusoidal function of the phase difference between θ_(ref),the phase of the voltage at the reference point 32, and θ_(inp), thephase of the voltage at the input port 37. Mathematically, this functionis:

    V.sub.out =a sin (θ.sub.ref -θ.sub.inp)+b

where "a" is device-dependent voltage scaling factor, and "b" is a fixedresidual offset voltage. For the purpose of demonstrating the functionof the system, the residual offset voltage can be assumed to be zero.Consequently, if the voltages at the input and output ports are eitherin phase (0° difference) or out of phase (180° difference), then thevoltage at the output port will be zero. Likewise for phase differencesof +90° and -90° the output voltages are +a and -a respectively. For allother phase differences, the output voltage varies between +a and -a ina sinusoidal fashion.

Phase Shifter--The phase shifter 30 is an insertable 2-port device. Withthe device energized at the input port 29, the phase of the RF voltageat the output port 31 can be changed by means of a signal applied to thecontrol port. The range over which the phase can be changed is 360°, andit can be changed either continuously, as in an analog phase shifter, orin a fixed number of steps, as in a digitally controlled phase shifter.For a fixed control signal, the phase change must not vary significantlyover the system operating band.

In this embodiment of the radar system, the phase shifter 30 isdigitally controlled. The phase of the signal at the output port can beset to any one of the 16 equally spaced 22.5° steps by means of a 4-bitdigital control signal.

D/A Selectable Voltage Source--The selectable voltage source 14sequentially applies one of four voltages to the VCO, each for a 0.02msec duration. For a single operating period, the sequence of fourvoltages is repeated 16 times for a total duration of 1.28 msec. Thetiming is controlled by the system synchronizer/controller 40.

Phase Shifter Controller--The phase shifter controller 46 applies a4-bit digital signal to the phase shifter 30. Each of the possible 16phase steps lasts for a period of 0.08 msec, synchronized with the A/Dselectable voltage source by means of the systemsynchronizer/controller. The total time interval required for all 16phase states is 1.28 msec.

A/D Converter/Sampler--The A/D converter/sampler 50 samples the voltageat the output port of the mixer 26 every 0.02 msec and converts thesampled voltage into a 12-bit digital number (-2046 to +2045). A totalof 64 samples is taken during the 1.28 msec period. The sampled data arestored in a digital form in Random Access Memory (RAM) for furthersignal processing.

System Synchronizer/Controller--The system synchronizer/controller's 40function is to synchronize the three control components described above.A total of 64 data points is taken, which can be divided into 16sequences of 4 data points, each sequence associated with one of the 16phase states. The sequences can therefore be designated by their phasestates. (A master clock 42 sets the basic repetition rate of the voltagesource 14 with its clock signals divided down by counter 44, therepetition rate for the 16 phase states is set.)

The total acquisition time of the 64 data points is 1.28 msec. In theautomotive application, where maximum relative speed between theantennas and an object does not exceed 12 inch/msec, any object can beconsidered essentially stationary during the acquisition period.

Focusing on the operation of two components, namely the power splitter16 and the mixer 26 (phase comparator), two distinct signal paths can betraced between them:

a) a reference path 28, defined by terminal points 27, 29, 31 and 32;and

b) an RF path, defined by terminal points 34, 33, (the target 22) 35 and37.

Let us assume that the RF path is 10 ft. longer than the reference path(the target 22 is thus at a distance of about 5 feet). In air, the RFenergy travels at the speed of 3×10⁸ meters/sec, equal to about 1ft/nsec. Thus, the time it takes the signal to travel through the RFpath is 10 nsec longer than the time it takes the signal to travelthrough the reference path. This additional time is referred to as thetime delay τ.

The RF signal can be described as a rotating vector, with one completerotation equivalent to one cycle. If the vector rotates only part of acycle, then that segment of the cycle can be described in terms of aphase change, where 360° corresponds to a complete cycle.

In similar fashion the VCO signal at any given frequency can bedescribed as a continuously rotating vector at a uniform speed, whoseangular velocity is determined by the frequency of the signal. As anexample, if the VCO is generating a signal at a frequency of 10.5 GHz,(10.5×10⁹) cycles/sec, or 10.5 cycles/nsec then the number ofrevolutions per nsec is 10.5.

At the beginning of each signal path, it is assumed that the referenceand the RF signals are in phase. In other words, their vectors at points27 and 34 are aligned and they are rotating at the rate of 10.5cycles/nsec.

Let us examine what happens to the alignment of the reference and RFvectors at the end of their respective paths (terminal points 32 and37). Since the RF signal takes 10 nsec longer to reach the end of thepath, its associated vector will have made 105 more revolutions than thereference signal (10.5 revolutions/nsec). Nevertheless, both vectorswill still have a zero net phase difference between them. Since themixer 26 acts as a phase comparator, the output signal of the mixer, atpoint IF, remains at its zero phase state for as long as the frequencyand the path length remain unchanged.

Now the case where the VCO frequency has been increased by 12.5 MHz,i.e., 10.5125 cycles/nsec, is considered. Consequently, in the same 10nsec period there are 105.125 more cycles of the RF signal vector.Therefore, at points 32 and 37, the vectors are 1/8 cycle or 45° apart,and the vector of the output signal of the mixer moves from its zerostate by 45°. Similarly, each time the VCO frequency is increased by12.5 MHz, the vector of the output signal of the mixer 26 is moved by45°.

In summary, when four frequencies are generated in the sequence f₀, f₀+12.5 MHz, f₀ +25 MHz and f₀ +37.5 MHz, the output vector moves throughan angular range of 135° in 45° increments (3/8 of a cycle in 1/8 cycleincrements). It should be evident that the angular range through whichthe output vector swings depends on the difference between the maximumand minimum frequency of the VCO. (The number of steps in which the VCOchanges from minimum to maximum frequency does not affect the angularrange of the output vector, except for a small quantization errorrelated to the frequency increment).

Based on the foregoing, the signal at the output of the mixer 26 can beanalyzed in terms of the information it may contain about the target.The angular range over which the output vector moves depends upon thedelay of the RF signal (hence the target distance). Thus, by measuringthe angular range of the output vector, the target distance can bededuced.

However, a problem arises when two or more targets are present. For eachtarget we can define a separate RF path and an associated RF vector. TheRF vectors may have various amplitudes and will span different angularranges, depending on the related distances. Correspondingly, eachassociated output vector also may have various magnitudes and willrotate at different speeds. The picture is now more complex. Sincevectors are actually superimposed on each other, the resultant vectorwill rotate at a non-uniform speed. In fact, it will exhibit a complexmotion, changing directions and amplitude, with uneven increments, eachtime the VCO frequency is increased by 12.5 MHz.

If, however, each vector by itself traces out one or more full cyclesduring the VCO frequency sweep, it is then possible to "decompose" theresultant vector into individual component vectors. This decompositionprocedure is known as "Fourier transformation". Provided the abovecondition is satisfied, we can obtain from the "trace" of the compositevector, via a Fourier transformation, the number of full revolutions foreach of the vector components. The locations of corresponding targetscan be readily deduced from the number of full revolutions of the vectorcomponents. It should be evident that this condition applies only totargets to 10', 20', 30', etc. For any other target locations, theindividual output vectors do not complete an integral number of fullcycles during the VCO sweeps, and, as it turns out, their distancecannot be resolved without an ambiguity.

This condition leads to a definition of the distance resolution.Resolution is defined by a distance needed to separate two targets, suchthat during the time the VCO frequency changes from minimum to maximum,the output vector associated with the further target will rotate onemore cycle than the vector associated with the closer target. Thiscondition is defined mathematically as:

    Δr=c/{2(f.sub.max -f.sub.min)}

where "c" is the propagation velocity.

As discussed above, for a case with 4 frequencies incremented by 12.5MHz and a target at 5 feet (corresponding to time delay of 10 nsec), theangular excursion of the output vector is 3/8 of a cycle.

The output vector will move through the full cycle, if we increase thefrequency bandwidth of the VCO by factor of 2. However, this is not adesirable solution.

The basic problem to be resolved is how to "force" the output vector tocomplete this cycle, so that as we repeat the sequence of VCOfrequencies indefinitely, the output vector rotates in a uniform andcontinuous manner as the VCO continues to sweep.

Let us examine what happens when the phase shifter 30 changes the phasein the reference path. For example, if that phase change is 180°, thenthe phase difference between the RF and reference signals at the mixer26, signals at points 32 and 37, changes by 180° and the position of theoutput vector shifts by a 1/2 cycle. A phase-shift occurs at each of theVCO frequencies. Correspondingly, the angular range of the output vectorwill again be 3/8 of the cycle, but beginning at the 1/2 cycle point.See FIG. 2 for a graphical illustration.

Therefore, as we continuously repeat the VCO sequence of 4 frequenciesand change the phase by 180° during alternate segments, the outputvector will rotate uniformly and continuously.

Since we have the required condition of a uniformly and continuouslyrotating vector, the principle of vector decomposition, i.e., theFourier transformation, can be applied to deduce the target distanceeven in the presence of other targets.

To summarize this new case, let us define a phase sequence as twosuccessive frequency sweeps. There are two possible phase sequencearrangements. (There will be a total of 8 data points, each associatedwith a given phase and frequency.) In the first arrangement, the phaseremains zero in both frequency sweeps, while in the second the phaseincreases by 180°. In the former instance, only the targets at 10', 20',30', etc. cause the associated output vectors to rotate 2, 4, 6, etc.complete revolutions per phase sequence. Analogously in the secondinstance, only the targets at 5', 15', 25' etc. cause the associatedoutput vectors to rotate 1, 3, 5, etc. complete revolutions per phasesequence.

Therefore, I have developed a method by which the apparatus is able toresolve a target with 5' accuracy, i.e., a factor of two improvement.The trade-off is that the apparatus needs to measure and analyze twiceas much data.

The above method can be extended for a larger number of phase steps,resulting in an additional improvement. In general, the improvementfactor is equal to the number of phase steps. If n_(p) is the number ofsteps, then the range accuracy is:

    Δr=c/{2n.sub.p (f.sub.max -f.sub.min)}

For an example, in the present embodiment the number off phase states is16. A phase sequence consists of 16 frequency sweeps. There are a totalof 16 phase states in the sequence arrangement. Let us designate each ofthem (0), (1), (2), . . . , (9), (A), (B), . . . , (F) (usinghexadecimal notation). For instance, in the sequence-0 the phaseincrement for each successive frequency sweep is zero, as the outputdata are measured. For sequences -1, -2 and -3 the increments are 22.5°,45° and 67.5°, respectively, and so on. Each of the corresponding datasequence must be analyzed for a different set of target locations, whichwill cause the output vectors to rotate one or more full revolutions persequence. Table 1 below demonstrates this example.

                  TABLE 1    ______________________________________    Sequence Phase     Targets with   No. of    designation             increment complete cycles                                      cycles    ______________________________________    -0       0° 0', 10', 20' etc.                                      0, 16, 32, etc.    -1       22.5°                       .625', 10.625', etc.                                      1, 17, 33, etc.    -2       45°                       1.25', 11.25', etc.                                      2, 18, 34, etc.    . . .    . . .     . . .          . . .    -4       90°                       2.5', 12.5', etc.                                      4, 20, 36, etc.    . . .    . . .     . . .          . . .    8        180°                       5', 15', etc.  8, 24, 40, etc.    . . .    . . .     . . .          . . .    -F       337.5°                       9.375', 19.375', etc.                                      15, 31, 47 etc.    ______________________________________

Digital processing

The fast Fourier transform (FFT) processes the data digitally. Themeasured data are samples of the signal produced at the output port 39of the mixer 26. One data sample is acquired for each frequency andphase state. There are a total of 4 frequencies and 16 phase states, or64 points, which are stored in a 1-dimensional array. DATA (I), I=0, . .. , 63. As can be deduced from Table 1 above, the phase sequence-1,generates all 64 combinations of frequencies and phases. Therefore, thissequence-1 is the only one performed during the measurements. The othersequences are then reconstructed in the computer processing. FIG. 4Ashows the flow chart which computes the frequency transformationresponse of FIG. 5. As each of the phase sequences is reconstructed, theFourier transform FFT is applied; only 4 points of the 64 FFT-outputpoints are required. These are indicated in the 4th column in Table 1above. These 4 points are stored in an array TABLE (k), k=0, . . . , 63(FIG. 4A). After 16 repetitions of the main loop, output array TABLE (k)is filled; a typical plot of TABLE (k) array is shown in FIG. 5, wherethe magnitude of each data point is shown.

The magnitude of the RF signal received at the antenna can have a largevariation, depending on the reflection of the target and its distance.For weak RF signals, random interference may be superimposed on themeasured data and the computed curve may not be as smooth as one shownin FIG. 5. It will have narrow spikes resulting in the "false" peaks. Itis therefore necessary to "smooth" the data stored in the array TABLE(k) of FIG. 4A by applying a smoothing algorithm, which is indicated inthe flow chart in FIG. 4B. The smoothing algorithm is based on widelyknown work (by R. B. Blackman and J. W. Turkey, "The Measurement ofPower Spectra", Dover, 1958) using FFT routines. Finally, a peak findingalgorithm is performed to determine index k_(max) at which the peak ofthe frequency transformation occurs. Finally, the target distance iscomputed; a suitable formula is set forth below (see Appendix, page 7,equation 16).

In a present embodiment of the system, the number of phase states n_(p)=16. For each of the phase states, the voltage controlled oscillator 12generates four frequencies f₀, f₀ +df, f_(O) +2df, f₀ +3df, where f₀=10.5 GHz and df=12.5 MHz. Each time a new frequency is generated, avoltage at the IF port 39 of the: mixer 26 is digitally sampled(measured) and stored in the computer's RAM memory; a total of 64measured data points are sampled and stored. FIGS. 3-A through 3-P are aset of 16 graphs (numbered 0 to F) each formed of the real portion ofthe 16 corresponding phase sequences developed in the reconstructionprocessing loop of the calculations process of FIG. 4A.

A typical example of a set of 64 data points is shown in FIG. 3-A. Thegraph is divided into 16 segments. Each segment represents a subset of 4data points (connected for clarity by a continuous curve).

Therefore, the first segment of the graph is a measured response at 4frequencies when the phase shifter 30 is in state 0°. The secondsequence is another response at the same 4 frequencies when the phaseshifter 30 is in state 22.5°.

Each subsequent segment corresponds to a response for 4 frequencies (f₀,f₀ +df, f₀ +2df and f₀ +3df), as the phase is increased by 22.5°. Thecomplete waveform is a set of 16 segments and is designated as phasesequence-1. The number of the sequence conveniently designates that inthis phase sequence, the phase state between subsequent segments isincreased by one least significant bit, LSB, (22.5°). The other 15 phasesequences, as shown in FIGS. 3-B through 3-P, are computer reconstructedfrom data contained in the phase sequence-1. Again, the phase sequence-0to -F refer to LSB increments of the phase shift between subsequentsegments.

The target range can be visually obtained by examining the 16 graphs forthe sinusoidaly continuous response and counting the number of fullsinusoidal cycles. In this example phase sequence-5 meets the criteriawith 5 complete cycles. This number of complete cycles directly relatesto the target range and is used for visual analysis of graphs. Machineprocessing via FFT (FIG. 4a) performs, a similar analysis.

A computer efficient method is to apply FFT to each of the 16 wave forms(phase sequences). FFT produces 64 complete data output points. FFToutput of phase sequence-1 produces data for 1, 17, and 33 etc.according to Table 1.

The resultant set of 64 points produces a curve as shown in FIG. 5. Theposition of the peak is related to the target distance.

The voltage controlled oscillator 12 can generate either a continuousrange or a discrete set of frequencies. In the former arrangement, thesystem can measure obstacle range from zero to infinity, being limitedonly by the strength of the echo signal.

The advantage of the latter case is that the system can be madesignificantly simpler and therefore cheaper. However, because of thefinite number of frequency steps, the range measurement has anunambiguity range:

    R.sub.u =c·n.sub.f /(2Δf.sub.m)

In the present embodiment, where the number of frequencies is four andthe bandwidth is 50 MHz, the unambiguous range is 40 feet. To overcomethis ambiguity problem, the system operates in two modes. In one of themodes, so called far-looking mode, the modulation bandwidth is 12.5 MHz,resulting in 160 feet of unambiguous range. It is reasonable to expectthat an echo signal from an obstacle past 160 feet will always benegligible compared to the signal received from an obstacle within theinitial 20 foot range.

The resolution in this far-looking mode is 2.5 feet. If a target isdetected within 20 feet, the system switches to the near-range highresolution mode, using a modulating bandwidth of 50 MHz with aresolution of 0.625 feet. In both of these modes the number ofgenerating frequencies is only four.

Single Antenna Implementation of Narrow Band Ranging Radar

A typical state-of-the-art monostatic radar system is depicted in FIG.11. Oscillator 113 generates a continuous wave RF signal with a periodicfrequency modulation. Power splitter 116, e.g., a Wilkenson powerdivider, divides the RF signal between output ports 127 and 134. Phaseshifter 130 is a 2-port device that can change the phase of the RFsignal. The phase shifter 130 provides an RF reference signal to a mixer126. The reference signal is applied to the LO port 132 of mixer 126.The RF signal is radiated by an antenna 120, however a portion of the RFsignal is reflected back from the antenna 120. Energy (v₂) reflected bythe target 122 and intercepted by the antenna 120 is fed to the RF port137 of the mixer 126. A circulator 158 ensures a maximum coupling of thetransmitted energy from oscillator 113 to the antenna 120 and of thereceived energy from the antenna 120 to the RF port 137 of the mixer126, and a minimum coupling (isolation) between the oscillator 113 andthe mixer 126. An alternative arrangement may employ a coupler insteadof the circulator 158. Receiver or detector 149 determines the range(ΔR) to the target 122.

In analyzing the monostatic system of FIG. 11, consideration must begiven to the reflection from the target 122 and the interaction with theantenna reflection (V₁). The signal voltage at RF port 137 equals thecombined reflection voltage from the antenna and the target.

    v.sub.RF =v.sub.1 +v.sub.2

In practice, the situation is more complex since a number of sources ofreflection and leakage energy are present. However, the solution to asingle false (undesirable) reflection case can be easily extended to thecase of multiple reflections.

Using conventional FM-CW techniques, the location of the target can bedetermined, provided that either, (a) the time delay difference betweenthe antenna.120 and target reflection is greater than the inverse of theeffective bandwidth, i.e., ##EQU1## where Δτ=time delay, c=the speed oflight, ΔR=range or the distance from the antenna to the target, andB=the effective bandwidth; or, (b) the RF voltage at the RF port 137 ofthe mixer 126 due to target reflection is significantly larger than thatof the antenna reflection, i.e.,

    v.sub.2 v.sub.1

where v₁ =the antenna reflection signal, and V₂ =the target reflectionsignal. Using the narrow band phase shifter technique requires only that

    v.sub.2 >v.sub.1

A typical microwave antenna reflects between 1% and 10% of the voltageincident upon it, while the RF voltage due to the target reflection, atthe RF port 137 of the mixer is anywhere between <0.01% to 1%. Clearly,this does not satisfy the preceding inequality.

One embodiment of the present invention is shown in FIG. 12. Thecorresponding components of the monostatic radar system shown in FIG. 12have the same reference numerals as the components in FIG. 11. A coupler160 is utilized instead of circulator 158. A one-port I/Q modulator orZ-network 162 is connected to one port of the coupler 160. The Z-network162 is an active matching device. The Z-network 162 "generates" avoltage v₁ ' at the RF port 137 of the mixer 126 that has an equalmagnitude and opposite sign to that of the antenna reflection voltagev₁. Since RF voltage is characterized as a vector quantity, theZ-network must be capable of both amplitude and phase control of thereflected signal. Generally, the antenna reflection will be a functionof frequency and temperature; therefore, it is preferred that theZ-network be adjusted accordingly.

One embodiment of the Z-network is shown in FIG. 13. The Z-network 162is composed of a bias circuit 164, a power divider 166, a 45°-phaseshifter 168, diodes 170, 172, and bias circuits 174, 176. A one-port I/Qmodulator is by definition a reflective device, and is characterized bya complex reflection coefficient Γ. The Z-network 162 "receives" theincident signal from the oscillator 113 via the coupler 160 and reflectsa pre-determined portion of that incident signal with a pre-determinedphase change. Ideally, the range of the amplitude of the reflectedsignal should be between 0 and 1, and the range of the phase should bebetween 0° and 360°.

Bias circuit 164 provides DC ground for the bias currents. Power divider166 splits the incident signal into two parts. A quadrature coupler canbe substituted for the power divider 166. One of the split signals mustgo through a 90° phase shift. This can be done by introducing a45°-phase shifter 168 into one of the paths. The phase of the signal isshifted 45° during each traversal of the path.

Each part of the split signal is reflected from a variable resistivetermination. The variable resistive termination can be properly biasedPIN diodes 170, 172 or GaAs FETs or other similar devices. The magnitudeof the reflected signals Γ₁, Γ₂ is determined by the resistance of thetermination. Nominally, the phase of the reflection will be a functionof resistance except for the 180° reversal when the resistance of thetermination moves through a value equal to the characteristic impedanceof the system. The reflections are recombined in a quadrature manner(because of the two-way transversal of the 45°-phase shifter 168 in onepath) to provide an I-Q relation between the two independentlycontrolled reflections, i.e.,

    Γ=Γ.sub.1 +jΓ.sub.2

Diodes 170, 172 are appropriately biased at each operating frequency andat each phase state of the phase shifter, if necessary, by bias circuits174, 176, respectively. The proper bias conditions are determined byminimizing the IF signal at IF port 139 in the absence of a target.

Since the diode resistance is highly temperature sensitive, thereflection coefficient Γ in FIG. 13 will also be temperature dependent.This is an undesirable condition. The Z-network 162 shown in FIG. 14asignificantly reduces the temperature dependence of Γ. Bias circuit 180removes unwanted RF interference. Power divider 182 evenly splits theincident signal between two paths. A 45°-phase shifter 184 provides thenecessary 90° phase shift to one of the split signals. Parasiticconditions and temperature can be compensated for by introducing aquadrature coupler and an additional variable termination in each path.The product of the resistances of diodes 188 and 190 must equal thesquare of the characteristic impedance of the quadrature coupler 186.Similarly, the product of the resistances of diodes 198 and 200 mustequal the square of the characteristic impedance of quadrature coupler196. This condition is achieved by applying the following bias voltages:

    E.sub.1 =E.sub.0 +V.sub.1

     E.sub.1 =E.sub.0 -V.sub.1

    E.sub.2 =E.sub.0 +V.sub.2

     E.sub.2 =E.sub.0 -V.sub.2

A voltage generating circuit, shown in FIG. 14b, consisting of a currentgenerator 208 and PIN diode 210 produces voltage E₀ (the voltage acrossdiode 210 biased by a current I₀). Current generator 208 has a T₀temperature dependance. ##EQU2##

Where T is the ambient temperature or system temperature in °K., and T₀is room temperature in °K.

V₁ is a DC control voltage with T/T₀ temperature dependance whichcontrols the magnitude and sign of the reflection Γ₁ via the followingtemperature independent relationship:

    Γ.sub.1 =a Tan h(bV.sub.1)

where a and b are physical constants. A similar relationship holds forV₂. In the preferred embodiment, diodes 188, 190, 198, 200, and 210should be lot matched. The article Broadband Phase Invariant Attenuator,Adler, D. and Maritato, P., delivered at the IEEE InternationalMicrowave Symposium, 6/1988, provides a complete description of thebiasing circuit; this article is herein incorporated by reference inthis application.

Temperature dependence of transmission line phase velocity must also betaken into account. However, its effect is not as pronounced as that ofthe diodes' resistance. The transmission line temperature variation canbe absorbed into the temperature variation of the antenna reflectioncoefficient.

The Z-network 162 of FIG. 14a is calibrated in the absence of anytarget, for each phase state i_(p) and frequency step i_(f). The DCvoltages V₁ [i_(p), i_(f) ] and V₂ [i_(p), i_(f) ] are determined byrequiring the IF voltage at port 139 lie within a predetermined range.For n_(p) and n_(f) steps there are a total of n_(p) ×n_(f) voltages ofV₁ and V₂. This procedure compensates for frequency variations of theantenna and phase shifter interaction at temperature T₀.

Over the temperature range, the temperature dependance of current I₀maintains the IF voltage within the predetermined range for all voltagesV₁ [i_(p), i_(f) ] and V₂ [i_(p), i_(f) ] obtained at temperature T₀.

An alternative monostatic radar system is shown in FIG. 15. This systemis a balanced monostatic system since the impedance is symmetrical withrespect to the Z-network and the antenna. The components of thisbalanced monostatic system are similar in function to those in FIG. 12.A 3dB quadratic coupler 222, with the isolated port terminated, is usedto equally divide the RF signal between the antenna 224 and theZ-network 228. This system employs a reflective phase shifter 226. Thereference LO signal is directed from port 291 via a coupling structure,including couplers 230, 232, 234 and 236, to a one-port phase shifter226 and its phase controlled reflection, v_(LO), appears at port 292 asan LO bias signal. The nature of the topology shown eliminates the LOsignal at port 293.

The RF signal is not phase controlled by the phase shifter 226. Thereare three components of the RF signal. A component v₁ ' is reflectedfrom Z-network 228; v₁ is due to antenna reflection; and v₂ is due totarget reflection. By calibrating the Z-network such that v₁ =v₁ ', thedetector 238 (i.e., a single ended mixer) at port 292 detects a signaldue to the target only. The antenna reflection v₁ is "dumped" in thetermination of port 293.

The significance of this approach is that the whole network structure issymmetrical, and, as a result, the temperature variation of thetransmission phase velocity is cancelled. If the antenna reflection istemperature insensitive, as in the case of a waveguide horn, then thereis no need for temperature compensation.

There are several shortcomings in this approach. A low-loss one-portphase shifter is not easy to realize. A four-bit phase shifter yielding16 fold bandwidth reduction is the maximum practical number of bits. Forhigher resolution, a higher loss I/Q network must be employed which willlimit the dynamic range of the system.

The use of single-ended mixer is another disadvantage. Such a mixer isless efficient than a double-balanced mixer used in FIG. 12. Inaddition, there is a 200-400 mV inherent DC offset. Therefore, an ACcoupling scheme should be employed.

FIGS. 16, 17, and 18 are particular realizations of the topology shownin FIG. 15. FIG. 16 is similar to FIG. 15 but the one port phase shifterand Z-network are realized by means of the one port I/Q network shown.The phase shifter is a balanced reflective phase shifter and includescouplers 256, 267 and 268 each with one port terminated; PIN diodes 258,260, 262, and 264; and 45° phase shifter 270. Note that the powerdivider is replaced by a quadrature coupler 267. The Z-network is alsobalanced and includes couplers 282,284 and 286 each with a portterminated; PIN diodes 274, 276, 278, and 280; and a 45° phase shifter272. FIG. 17 shows an unbalanced Z-network which is essentially theZ-network of FIG. 13 with the power divider replaced with a quadraturecoupler 314. A temperature compensated balanced Z-network is shown inFIG. 18, which is essentially the Z-network of FIG. 14a with the powerdivider replaced by a quadrature coupler 414. The Z-network alsoincludes couplers 416 and 418 with one port terminated; PIN diodes 426,428, 430 and 432; and 45° phase shifter 420.

FIG. 19 shows the monostatic radar system incorporating the presentinvention including a digitally controlled phase shifter 500 and theone-port active Z-network 502 (antenna active matching device).

The foregoing describes a new and improved near range obstacle detectionand ranging aid. The invention improves on existing technology toprovide a sufficiently high resolution of targets without requiring anincrease in bandwidth beyond what is presently available. In addition,multiple targets can be identified.

Applicant has disclosed various forms and modifications of the inventionand others will be apparent to those skilled in the art from theconcepts set forth above and in the following claims.

The attached Appendix sets forth theoretical bases for this inventionand is incorporated herein as a part of the application.

Appendix Narrow-band Radar System With Improved Range Resolution

Overview

The specific problem addressed in a particular automotive applicationhas been to measure the range of a target, (or the closest one in amulti-target environment), with less than one foot resolution over a 0to 20 feet range. Among the various approaches considered, a microwaveFM-CW radar has been chosen. It is a mature technology with low costattributes, adequately suited for the automotive environment and offersmany advantages compared to ultrasonic, infra-red or optical techniques.The principal limitation encountered in this approach stems from therestricted operating bandwidth allowed by the FCC for unlicensed radartransmission.¹ At the preferred operating band, nominally at 10.525 GHz,bandwidth is 50 MHz.

It is known that the inherent range resolution Δr, is directly relatedto the radar the allowable bandwidth Δf_(m), i.e. ##EQU3## where c isthe velocity of propagation, which for electromagnetic energyapproximately equals one ft/nsec in free space.² This is also referredto as the critical distance problem.³

Thus, for Δf_(m) =50 MHz bandwidth, the inherent resolution according toequation (1) is Δr≈10 feet.

In the most broad sense, the key aspect of an FM-CW system is themeasurement of a relative time delay between two coherentfrequency-modulated signals. Improved resolution can be achieved byphase modulating, over repeated cycles of frequency modulation, one ofthe two signals. In so doing, additional time is needed to acquirecomplete data under all phase conditions. This is an example of theclassic time/bandwidth trade-off.

A typical FM-CW system is shown in FIG. 20. The LO signal, having afrequency F_(c), is frequency modulated over a bandwidth Δf_(m).Assuming linearly swept modulation, the instantaneous frequency variesbetween F_(c) and F_(c) +Δf_(m) B, over the finite swept period Tstarting at some arbitrary time, as shown in eqn. (2), as follows:##EQU4##

The parameter of interest is the electrical phase of the oscillatorsignal θ_(LO), shown in equation (3) which includes as arbitrary phaseψ_(o). ##EQU5##

The oscillator signal is split into two parts, for example by means of apower divider as shown in FIG. 20. These signals are then guided througha reference path and a transmission path, respectively. Referring toFIG. 20, the former is the distance between the power splitter and theLO port of the mixer, while the latter is the distance between the powersplitter and the RF port of the mixer. In FIG. 20, the transmission pathincludes a distance 2r, the distance between the transmitting andreceiving antennas via the target.

Without the loss of generality, we can treat the problem by assumingthat all distances, except the distance 2r, are zero. Therefore itfollows that the phase of the reference signal at the LO port of themixer is also given by equation (3).

The transmitted part of the oscillator signal which travels a distance2r, is received with a delay τ=2r/c. Thus the phase of the signal at theRF port of the mixer, θ_(RF), is related to the phase of the signal atthe LO port by equation (4).

    θ.sub.RF (t)=θ.sub.LO (t-τ)                (4)

The function of the mixer in FIG. 20 can be characterized as phasecomparison. Thus, if the LO and RF signals are sinusoidal waveforms withconstant amplitude, then the waveform at the IF port is also asinusoidal function whose phase, θ_(IF), equals the difference betweenthe phases of the LO and RF signals, or ##EQU6## where ψ₁ is a timeindependent constant phase and the second term represents the IFfrequency term proportional to the delay τ=2r/c.

Some observations about the nature of the IF signal:

1. The IF signal is a periodic function.

2. Therefore all of the available data can be acquired over the periodT.

3. Evidently, in the single target case, the delay τ can be measureddirectly from the phase response, by determining the total phase changebetween the beginning and the end of the sweep period T. The measuredresolution would be limited only by the capability of a phasediscriminator employed. However, in a multiple target situation, thedirect phase measurement produces large errors.

4. If the product Δf_(m) τ is an integer, the IF output will be acontinuous sinewave. Otherwise it will be a repetitive discontinuoustrain of sinewave segments. In the former case, the output phase isswept over a full 2πn range. In the latter case, at least one phasecycle is incomplete. It is evident from equation (5) that the phaseconstant ψ₁ defines which part of the 2π cycle remains incomplete.

5. In the frequency domain, the IF signal, as a consequence of paragraph1, above, is a discrete function with its harmonic components being 1Tapart.

6. For each harmonic there is a corresponding discrete location of atarget with a delay τ_(k) =k/Δf_(m), where k=0, 1, 2 . . . is theappropriate harmonic number.

7. If k in above is not an integer, then the frequency domain responseto such a target will be two or more harmonics.

8. It follows, that the resolution is limited by the condition Δf_(m)τ=1, which confirms equation (1).

Modified FM-CW approach

The key aspect in the modified FM-CW system is the inclusion of avariable 2π phase-shifter in one of the two signal paths, steppingthrough n_(p) discrete phase states over repeating cycles of frequencymodulation. The phase of the phase-shifter ψ_(LO) (t) has a time varyingnature with a period n_(p) T, since the period of the frequencymodulation cycle is T. In FIG. 21, the phase-shifter is shown as beingin the reference path. It can be readily shown that the phase of the IFsignal is similar to that of equation (5), except it includes phaseψ_(LO) (t) of the phase-shifter, i.e. ##EQU7## For each sweep period T,the phase at the LO port is changed in n_(p) steps.

Several observations can be made:

1. The IF signal has a period of n_(p) T.

2. It follows that in the frequency domain the discrete frequencies areseparated by 1/n_(p) T.

3. Thus the necessary condition for improved resolution exists.

4. The range resolution is given by ##EQU8## For example, by using 4 bitphase-shifter, n_(p) =16. The resolution for Δf_(m) =50 MHz would be,according to equation (7), 0.625 ft.

Digital Signal Processing

In this approach the IF signal is sampled with n_(f) samples perfrequency sweep. Consequently the frequency modulation cycle of theoscillator can have n_(f) discrete frequencies. However, because of thefinite number of frequency steps, the range measurement has anunambiguity range R_(u) readily shown to be ##EQU9## With n_(p) phasestates per each frequency sweep, the total number of acquired data overa period n_(p) T is

    n=n.sub.p n.sub.f                                          (9)

Each of the n data has a unique pair of frequency and phase indexesi_(p), and i_(f). These data are designated as v(i_(p), i_(f)). Let usdefine a transformation of v(i_(p), i_(f)) as follows ##EQU10##v(i_(p),i_(f)) i_(p) =O,1, . . . , n_(p) -1 and i_(f) =0,1, . . . ,n_(f) -1 (n_(p) n_(f) total), is a set of acquired data points, definedto be periodic, equation (11).

    v(i.sub.p +mn.sub.p, i.sub.f)=v(i.sub.p, i.sub.f) for m=any integer(11)

It can be shown that, for a single target, the magnitude of thetransformation, shown in equation (10), leads to a sin(x)/x response.

An alternative form of equation (10) can be obtained by defining a newindex i, as follows:

    i =i.sub.p n.sub.f +i.sub.f                                (12)

Then equation (10) reduces to ##EQU11## Range Determination

Let k_(max) be an index number of the Z(k) transformation such that

    |z(k.sub.max)|=max|Z(k)| for 0≦k≦n-1                                     (14)

In other words, k_(max) is the index of the complex array Z(k), forwhich the magnitude of Z(k) is maximum. Then the time delay τ=2r/c isrelated to the index k_(max) by equation (15). ##EQU12## CableCorrection

In practice the distance with the reference path contains cable orsimilar transmitting media with a known time delay τ_(ref). Similarly,the transmission path includes cables between the power splitter andtransmitting antenna, as well as between the receiving antenna and theRF port of the mixer, having known time delays τ_(a) and τ_(b). Itfollows that ##EQU13## Leakage Correction

In a practical system one or more leakage paths may exist between the RFand LO ports of the mixer. When measuring a target with a weak echosignal, a stronger leakage signal may cause significant errors. Sincethe transformation of equation (13) has a commutative property, we cangenerate a corrected signal u_(corr) (i)=u(i)-u_(cal) (i), which is tobe used in equation (13). The signal u_(cal) (i) is measured when notargets are present. Alternatively, we can measure u_(cal) (i) even inthe presence of targets, if both antennas are replaced by a matchedload. In this case, however, the external leakage between the antennascannot be corrected and therefore will limit the useful dynamic range ofthe target echo.

Multiple Targets

Let us assume that a total of L targets is present. We have Ltransmitting paths of different lengths. Due to each transmitting paththere is an associated signal v_(l) (i) and transform the Z_(l) (k) foreach target 1=1,2, . . . L. It is readily seen from commutativeproperties of (10) or (13) that the combined effect of all targetsresults in transform Z(k) such that ##EQU14##

Since each transformation Z_(l) (k) is sin(x)/x function, the combinedtransform is a summation of L such function having maxima shifted withrespect to each other with varying magnitudes of the maxima. Someanalysis as well as experimental measurements indicate several aspects:

1. For targets which produce about equal signals at the receivingantenna and are separated by several units of Δr, the measured rangewill be a average of two target distances.

2. For targets which produce about equal signals at the receivingantenna and are separated by several units of Δr, the measured rangewill be a distance corresponding to the distance of the stronger target.

3. For targets which produce about equal signals at the receivingantenna and are separated by many units of Δr, the range of both targetscan be obtained from the measurements.

4. For targets which produce unequal signals at the receiving antennaand are separated by many units of Δr, the range of the stronger targetsis obtained accurately. The discrimination of the weaker target ispossible, provided the magnitude of its signal is within 10 dB of thestronger signal.

In a particular automotive environment, we are always interested in theclosest target. Since signal strength decreases 12 dB each time thetarget distance is doubled, we directly benefit from rangediscrimination of farther targets.

I claim:
 1. Range detection apparatus comprising:(a) means forgenerating high frequency energy over a finite frequency range and forfrequency modulating over a limited range of bandwidth said highfrequency energy and for supplying said frequency-modulated highfrequency energy to a first electrical path providing a phase reference,(b) a second electrical path for transmission and receiving andincluding means for radiating said frequency modulated high frequencyenergy into space in the form of propagating waves and for receiving aportion of said radiated energy after reflection from a remote object,said energy portion acquiring a phase shift related to the distancetraveled by said radiated energy and to the frequency of said radiatedenergy, (c) means for phase shifting said frequency-modulated highfrequency energy at a plurality of predetermined frequency values over afinite frequency range of said bandwidth and at a predeterminedplurality of phase-shifting values having a certain quantity greaterthan two, and at combinations of said phase-shifting and said frequencyvalues to improve by a factor of said certain quantity the relativerange resolution attainable with said limited range bandwidth, (d) meansfor comparing the phase of said frequency-modulation energy in one ofsaid paths with the phase of said reflected and received radiated energyin the other paths to produce range measurement signals related to thecombined phase shifts of said two paths and in accordance with saidsignals produced by said phase comparing means, and with thecombinations of phase states of said phase shifter and the frequenciesof said energy generating means, (e) means for deriving rangemeasurement information from all of said phase-shifting and saidfrequency values; and (f) said phase shifting means is in said secondpath.
 2. A method for detecting range comprising:generating highfrequency energy over a finite frequency range; frequency modulatingsaid high frequency energy over a selected frequency range to generate aplurality of predetermined frequencies; supplying saidfrequency-modulated high frequency energy to a first electrical pathproviding a phase reference signal; radiating said frequency-modulatedhigh-frequency energy into space in the form of propagating waves from asecond electrical path and receiving a portion of said radiated energyafter reflection from a remote object as a reflected received signal,said reflected received signal acquiring a phase shift related to thedistance traveled by said radiated energy and to the frequency of saidradiated energy, phase shifting said frequency-modulated high frequencyenergy in one of said paths at a predetermined plurality ofphase-shifting values for each predetermined frequency, comparing thephase of said phase reference signal with the phase of said reflectedreceived signal to produce range measurement information related to thephase and frequency of said signals.
 3. The method of claim 2, whereinsaid phase shifting is in said first path and operates to phase shiftsaid frequency-modulated high frequency energy supplied thereto.
 4. Themethod of claim 2, wherein said phase shifting is in said second pathand operates to phase shift frequency-modulated high frequency energysupplied thereto.
 5. The method of claim 2, wherein said step ofgenerating high frequency energy includes generating a continuous set offrequencies.
 6. The method of claim 2, wherein said step of generatinghigh frequency energy includes generating a set of discrete frequencies.7. The method of claim 2, wherein said step of producing rangemeasurement information from said signals comprises:comparing the phasesof said signals to provide an output vector signal; sampling anddigitizing said output vector signal; and storing the digitized signal;and processing said sampled, digitized and stored output vector signalin association with its corresponding phase and frequency to producerange measurement information.
 8. The method of claim 7, wherein saidstep of producing range measurement information further comprisestransforming said output vector signals from the time domain to thefrequency domain.
 9. The method of claim 8, wherein said step ofproducing range measurement information further includes generating asignal distribution of power spectrum density for certain phase andfrequency combinations and producing range measurement information fromthe maximum of the distribution.